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    • 专利摘要:
    The subject invention describes a method for controlling the torque of the asynchronous machine in the stator-fixed coordinate system without having to use a known flow model based on a voltage and current model. This is achieved by the stator flux step (d S (n + 1) to be executed in the next sampling step (n + 1) in stator-fixed coordinates (a, β) starting from the current rotor flux (R (n)) and stator flux (S (n)) )) to achieve a desired torque, and the rotor flux (R (n + 1)) and stator flux (S (n + 1)) for the next sampling step (n + 1) are calculated.
    公开号:AT511134A2
    申请号:T50200/2012
    申请日:2012-05-24
    公开日:2012-09-15
    发明作者:
    申请人:Voith Turbo Kg；
    IPC主号:
    专利说明:

    J
    Printed: 25-05-2012 E014.1 ^ 10 2012/50200
    VT-3478 AT
    Method for controlling the torque of an asynchronous machine
    The subject invention relates to a method for controlling the torque of an asynchronous machine.
    Above an inverter driven asynchronous motors are often regulated according to the principle 5 of field-oriented control. The basic idea of the field-oriented control is known to be to consider the electric machine in a co-rotating with the magnetic flux coordinate system, which is based on the magnetic flux. The stator current may then be decomposed into a field-forming current component in the direction of the flux, and a torque-forming current component normalized to the direction of the flux. From the setpoint values for torque and magnetic flux, the corresponding setpoint values for the torque-forming and field-forming current components are determined. In the simplest case, the voltage space vector to be set in the field-fixed coordinate system is determined with one PI controller for each current component. By means of a coordinate transformation, the voltage space vector to be set is calculated in the stator-15 fixed coordinate system and, using a suitable pulse modulation method, the drive signals for the three half-bridges of the inverter are determined therefrom.
    In addition to the calculation of the field-forming and torque-forming desired current components, the determination of the current flux in the asynchronous machine is of central importance in the implementation of the field-oriented control. Since the flow is not accessible to a direct, robust and insensitive measurement, a so-called flow model is used to calculate the flow or to observe the flow from the measured currents and the voltage applied to the motor.
    With regard to the concrete implementation, there are different ways of implementing the field-oriented regulation. In principle, both the field orientation at the rotor flux and at the stator flux is possible. For the former variant, simpler equations result, so that often the field orientation at the rotor flux is preferred.
    From EP 674 381 A1 a method for controlling the torque of an asynchronous machine is known, which is based on the stator flux and with a pulse width modulation leads the stator flux on a circular path. The method is based on a flow model that observes the stator and rotor flux from the measured currents and the applied voltage in the stator fixed coordinate system. The regulation is based on the stator flux and determines a voltage component pointing in the direction of the stator flux and a voltage component normal thereto. In the simplest version, the voltage component pointing in the direction of the stator flux is determined directly by the output of a PI controller acting as a field controller, to which the control deviation between the current '1' 24-05-2012
    Printed: 25-05-2012 E014.1
    10 2012/50200 VT-3478 AT the stator flux amount and its setpoint is supplied. The normal voltage component is determined directly by the output of another PI controller acting as a torque controller to which the control deviation between the current rotor circuit frequency and its setpoint is applied at the input. The thus determined stator voltage space vector is 5 transformed into the statorfeste coordinate system and impressed with a pulse width modulation method at the terminals of the machine. By pre-controlling the controller outputs, the PI controllers are relieved statically and dynamically, so that finally they only have the task of correcting remaining errors in the feedforward control. The rotor frequency setpoint is limited in such a way that, on the one hand, the stator current remains limited to permissible values (current limitation) and, on the other hand, the asynchronous machine does not tip over (protection against tilting).
    German Offenlegungsschrift DE 103 36 068 A1 discloses a field-oriented method for controlled impressing of the stator current and torque of a converter-fed induction machine. In this case, the field and torque-forming current components from the torque setpoint and the rotor flux setpoint and the stator circuit frequency are determined in a coordinate system oriented on the rotor flux, wherein a flow model determines the required rotor flux actual value. The current components and the machine parameters are used instead of the stator voltage space vector to calculate a so-called terminal flux setpoint, which specifies the voltage time area to be impressed by the inverter (ie the integral of the terminal voltage). In the pulse pattern generator, the flux path curve predetermined by the terminal flux setpoint is now impressed into the machine with the aid of off-line optimized flow path curves. This gives an instantaneous value-oriented control of the position of the stator current to the rotor flux, whereby the motor currents and the torque of the induction machine are controlled stationary and dynamic exactly 25. The method combines the high control dynamics of directly switching processes with the optimal stationary behavior of off-line optimized pulse patterns.
    Common to the known methods is that they use a flow model consisting of a voltage and a current model to determine the position of the rotor flux, which calculates the rotor and stator flux from the measured currents and the voltage impressed by the inverter, which is generally takes place in a stator-fixed coordinate system. The voltage or voltage time surface to be impressed by the inverter is calculated by controllers in fixed-field-fixed coordinates. These methods are based on feedback control, whereby the governors can be precontrolled for dynamic unloading based on inverse machine models. Coordinate transformations between the rotating field and the stator-fixed system are therefore required for both the measured current space vector and the manipulated variable vector.
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    VT-3478 AT needed. Although feedback regulations have the advantage that errors in the machine model or in the parameterization of the machine model are compensated by the control and thus hidden. At the same time, however, model errors are very difficult to identify. On this basis, it is an object of the subject invention to provide a method for controlling the torque of the asynchronous machine in the stator-fixed coordinate system, without using the known based on the voltage and current model flow model. Nevertheless, a high torque dynamics and accuracy of the torque control should be achievable. This object is achieved according to the invention by the stator flux step (δψ8.) To be executed in the next sampling step (n + 1) in stator-fixed coordinates (α, β) starting from the current rotor flux (ψκ (/ 7Γα)) and stator flux ( | / s)) (nTg + Ta)) to obtain a target torque, and the rotor flux (ψκ («Ζ], + Ta)) and stator flux (ys (nTa + Tä)) for the next sampling step (n + 1) is calculated. This results in a feedforward control in fixed-stator coordinates, which makes it possible to dispense with a separate flow model for determining the rotor flux. In this way, a highly dynamically controlled pilot control of the asynchronous machine, which follows in the torque dynamics directly to the desired value. This also makes it possible to control the asynchronous machine solely on the basis of the state variables (of the flow), and above all without having to explicitly calculate the currents and voltages on the machine. Due to the reference to the stator-fixed coordinate system, no differentiation of the flow quantities occurs in the machine equations. Differentiation can only ever be calculated approximately and tends to amplify measurement noise. Furthermore, this eliminates the coordinate transformations of the manipulated variable between the rotating field-fixed and stator-25 fixed coordinate system and the determined Statorflussschritt in statorfesten coordinate system can be converted directly into the control commands of the inverter without having to calculate a voltage space pointer.
    The tracking of the parameters of the machine model, which is usually necessary in a feedforward control, can be done easily, since the parameters change only 30 slowly and therefore need not be updated with high frequency, and thus low computer requirements.
    Finally, an as precise as possible machine model of the asynchronous machine which is to be preferred for the feedforward control and which directly reproduces the physical behavior, is a very highly appreciable advantage, since it makes it possible to obtain further parameters.
    -3- 124-05-2012
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    VT-3478 AT ßen derive from the machine model and not to measure. In this way, e.g. to dispense with certain sensors.
    Further advantageous embodiments and advantages of the invention will become apparent from the dependent claims, as well as the following description of the invention. The present invention will be explained in more detail below with reference to FIGS. 1 to 4, which show exemplary and schematically advantageous embodiments of the invention. It shows
    1 is a schematic diagram of a traction drive of a rail vehicle with an asynchronous machine, io Fig.2 the control model underlying an asynchronous machine model,
    3 shows a block diagram of the control according to the invention and FIG. 4 shows a block diagram of a current-limiting device and a tilt-limiting device.
    FIG. 1 shows by way of example a traction drive of a rail vehicle. The three-phase asynchronous motor 8 of the rail vehicle is driven via a traction converter 1 with a voltage source inverter. In this case, the intermediate voltage circuit 6 of the traction converter 1 is connected via a feed circuit 5 to the tapped with the current collector 3, applied between the overhead line 2 and the rail 4 voltage. If necessary, the feed circuit 5 converts the input voltage into a suitable DC voltage, which feeds the inverter 7 at the voltage intermediate circuit 6. The aim here is to drive the inverter 7 of the traction converter 1 with a control device 9 so that a rotational voltage variable amplitude and frequency is generated at the terminals of the induction motor 8 in such a way that the induction motor 8 on its shaft 10, the desired torque both static as well as 25 dynamically generated. In this case, the limitations of the inverter 7 in terms of the maximum adjustable output voltage, the permissible output current and the permissible switching frequency are normally observed, and to keep the output current with regard to the losses in inverter 7 and asynchronous 6 as small as possible. The control device 9 is the rotational speed of the drive 30 detected by a rotary encoder 11 as well as the motor currents detected by a measuring device 12 as actual values.
    The subject invention is of course not limited to traction drives, but can generally be used for the control of an asynchronous motor.
    In known field-oriented control methods, the asynchronous machine is described in a rotating field-fixed coordinate system running with the stator field, and the
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    10 2012/50200 VT-3478 AT
    Stator current impressed by the control. Therefore, an equivalent circuit diagram of the asynchronous machine 8 is preferably used, in which the leakage inductance is completely added to the stator. In contrast, asynchronous machine 8 is described herein by equations in a fixed-stator coordinate system (a, β), e.g. based on the equivalent circuit shown in Fig.2 5. Here, the leakage inductance is completely added to the rotor and the iron losses are taken into account by an arranged parallel to the Statorinduktivät ohmic resistance. As can be seen from FIG. 2, the machine model is divided into two submodels, wherein the "inner" submodel 200 "behind the stator resistance" can be understood as a model of an idealized machine without stator resistance and without iron losses. The "outer" submodel 201 forms the
    Stator copper losses and iron losses. This model approach makes it possible to account for the iron losses in a relatively simple manner, while the remaining equations of the "inner" submodel remain unchanged compared to a machine model without consideration of the iron losses. Based on this machine model, the dynamics 15 of the asynchronous machine 8 is described by the following equations
    Stator bag: /. , <ILs = ^ + ^) + 7 'a τ Eq. (1) Rotor pocket: άψ - ~ Γίϊ-Λ ~ R Μη, Ψ »Eq. (2) stator linkage: = + Lr) Eq. (3) Rotor flux linkage: ΨΗ =! SLs &gt; + {! s' LY »Eq. (4) Torque: Eq. (5) where the individual quantities are given the following meaning: complex stator current space pointer complex stator current space vector of the inner submodel 4 complex stator current space vector of the inner submodel [R complex rotor current space vector tFe complex current space vector replicating iron losses Ίί * complex stator flux linkage space vector complex rotor flux linkage space vector rs stator resistance rR Rotor resistance! S stator inductance / σ leakage inductance concentrated on the rotor side τ normalized time (om mechanical angular frequency of the asynchronous machine -5-
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    10 2012/50200 VT-3478 AT j imaginary unit - / =
    By transformation, a differential equation for the flux linkage ψ R is obtained from these equations, with Αψ = ψ s -ψ R for the leakage flux linkage. d r = j (o ψ + ~ Δ ψ di J m ~ R 1 -
    From this it can be seen that leakage flux linkage Αψ can be used directly as a manipulated variable for
    Control of the rotor flux can be used. The rotor flux in the asynchronous machine is accordingly controlled by a direction in the direction of the rotor flux, e.g. given by, oriented - i. field-oriented specification - the leakage flux linkage Αψ_ = [Αψχ +] Αψ}) eVs, Eq. (7) 10 controllable, wherein the pointing in the direction of the rotor flux stray flux component Αψχ field-forming and the normal to the rotor flux stray flux component Αψγ has a torque-forming effect.
    If we write instead of the complex space vector ψ = ψΚα +] ψΆρ and Δ ψ - Ay / U +] Αψρ the rotor flux vector (state vector) ψκ = and the scattering flux vector (manipulated variable vector) Δψ = (δ «//" Ay / ßJ in stator-fixed Coordinates (a, ß), then follows 15
    Eq. (6)
    Ψ the unit vector e, "= ~ R from the elementary Eq. (6) directly the time-continuous state space representation of the rotor flux integrator in the stator-fixed coordinates (a, β): - VR = AcvR + BtAv Eq. (8) ατ with the system matrix Ac = 0 -G). ^ '1 &lt; Λ and the control matrix - -, The Ansteuer- 0 K j tion of the inverter is to be done with a pulse pattern that imprints the leakage flux vector (manipulated variable vector) Δψ in time discretized sampling steps nTa, with the sampling time Ta in the machine. From the time-continuous state space representation, therefore, an equivalent time-discrete state space representation is computed, which according to ΨΨ ("7 &0 = A ^R (77;) + BdAV &7;+7;) Eq. (9) νω> »
    Eq. (9) 20 5 VT-3478 ΑΤ the rotor flux vector ψΗ (/ ίΓα + 7 ^) at the end of the next sampling step (n + 1) from the rotor flux vector ψκ («7α) of the current sampling step (n) and in the next sampling interval (n +1) to be applied leakage flux vector (manipulated variable vector) Δψ_ (nTa + Ta) calculated.
    The notations (n) and (nTa), as well as (n + 1) and (nTa + Ta) are to be regarded as the same. The minus in the index indicates that in Eq. (9) the manipulated variable vector Δψ is to be used in its orientation at the beginning of the sampling interval, that is to say at the sampling instant nTa. The calculation of the time-discrete system matrix Ad and control matrix Bd is a standard control technical problem that is not explained in detail here. For the time-discretized system matrix Ad = cos-sincvr y, one obtains a rotation matrix that rotates vSin o> Yes the rotor flux vector by the angle ojmTa, e.g. is detected with the encoder 11. This corresponds to the mechanical rotation of the rotor during the sampling interval. With the course of the manipulated variable vector during the sampling period (constant amount with uniform rotational movement about the angle cosTa, with the stator circular frequency ως) the time discretized control matrix is calculated to ** = τ.γ sin ^ cos sin - 2 (ο. + Ω "T , - sin »s + ω» Λ cos 2
    The leakage flux vector (manipulated variable vector) performs a uniform rotational movement Δψ (τ) during the sampling interval.
    vsinß) jT
    - sin cosT COSOPT Αψ. (»7; + 7;) with y
    Eq. (10) t £ [0 T0 and has rotated by the angle ojsTa until the end of the sampling interval (τ = Ta). Therefore applies to the manipulated variable vector Δψ. at the end of the next sampling interval
    Eq. (11) Δψ + {nTa + Ta) = ΑΑΔψ. {r Ta + Ύα) with the rotation matrix AÄ rcos cosTä vsin tosTa -sin 0,7 ^ cosos7; j
    From the leakage flux linkage Αψ = ψ s -y / and Eq. (11), the stator flux vector ys {ni'a + Ta) is calculated to be + (nTa + Ta) * GI (12) -7-20 at the end of the next sampling interval
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    10 2012/50200 VT-3478 AT During the next sampling interval, the stator flux vector + Γα must therefore be
    Statorflussschritt
    Sys (nTa + Ta) = ψ5 {nTo + Ta) ψ5 {nTa) Eq. (13) in order to impress the desired leakage flux vector (manipulated variable vector) Αψ in the asynchronous machine 8. 5 From the equations Eq. (13), Eq. (12) and Eq. (9) therefore results directly, as in stator-fixed coordinates (α, β), from the current rotor flux yR (nTa) in the next
    Sampling step (nTa + Ta) to be performed stator flux step 6ys (nTa + ΤΊ) can be calculated.
    J-S
    From the stator voltage equation Eq. (1) it can be seen how the stator flux linkage ψ 10 can be influenced by the voltage space indicator us applied to the machine terminals. By integrating this equation over the sampling interval Ta one obtains -t Ϊ '
    VjPTa + Ta) = f "S ^ T - JrAs, + (/, - + ^ (, ϊ7") nT ° "T °, from which for the stator flux step δψ8 (/ Γo + Ta) of the stator flux vector in vector notation nTa .Ta "T, <r" δψκ ("^ + ^) = ψ8 (" 7> 7ρ) -ψ8 ("7α) = frvOsi + 'FeV ^" T ° "T * follows. The requested 15 stator flux step ö * | / s (wTa + Ta) can thus be impressed by a pulse pattern generator when the inverter 7 at the motor terminals in the next sampling interval (nTa + Ta) ιιΊ '+ Τ "a voltage time surface Jusi / r = δψ8 (/ ίΓα + 7ο) + JYs (iSj + ΐΡβ) Λ ·, which is equal to the target stator flux step δψ8 (nTa + Ta) plus the integral of the ohmic stator voltage drop over the sampling interval Stator voltage space pointer us match
    Switching interval of duration Ta known. In principle, these methods are also suitable for realizing a stator-flux-influencing actuator when the mean stator voltage space vector from the stator flux step δψ5 {ηΤβ + Τσ) becomes us = - Jusi / r = - ^ δψ8 ("^ + 7" ") + r5is with the average stator current space vector '' Ϊ '/' -8-
    Printed: 25-05-2012 E014.1
    10 2012/50200 VT-3478 AT nT "-n_ ι / s = 77- J /> (iSi + iFe) ir. The Statorstromraumzeiger the inner submodel 7 a ηΐ, behind the stator resistance can then with the Flußverkettungsgleichungen Gl. (3) and Eq. (4) from the flux linkage space pointers to 4+ '«1 77 ~ ψ5 ~ ΓΨκ Vcr
    Eq. (14). 5 The specification of suitable stray flux components oriented on the rotor flux (Δι // Χ, Δψν), see Eq. (7), results from determining the field-forming leakage flux component Δψχ by a proportionally acting flux controller from Δι //, with ψΗ =, ΙψϊϊΤψ ^. The proportional gain kp of the flow controller is known to control the flow control dynamics (the higher kp, the faster the desired flow value is set), the required manipulated variable (the higher kp, the greater the required stator voltage step for flux deviations) and the susceptibility to interference (if the Rotor flux amount "noises", so at large kp this noise is transferred to the manipulated variable). In summary, as with any proportional regulator, kp is set as a compromise between good control dynamics and the required interference suppression. The torque-forming leakage flux component Δψν is determined from
    Moment equation Eq. (5) determined. If you press in Eq. (5) the stator current according to Eq. (14) through the flux linkages, the torque results after a short calculation to w »= -v /) = - If we now consider ψ y = ψ + Δψ, one obtains with Δψ = Δψχ +] Δψ}, in spin-field-fixed stator coordinates (x, y), in which, by definition, = | i // fi and = 0, the torque becomes m = - Δψ}, ψR. To a desired
    To generate torque mSon, therefore, the torque-forming component Δψ}, the leakage flux to Δψ, = KmSn, l 1 Ψ *
    Eq. (15). In addition, it is possible to calculate the stator circuit frequency ujs to 'setpoint.
    ψ Z-R -9- E014.1
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    VT-3478 AT
    This makes it possible to realize a feedforward control of the torque m of the asynchronous machine 8, as will be described below with reference to FIG. 3 in an advantageous embodiment.
    A flow controller 301 calculates nTa from the deviation of the rotor flux amount | ψη | determined by a flux integrator 306 in accordance with the above statements for the current sampling step 5 from the predetermined desired value | ν / Λ | amp / the field-forming leakage flux component Δψχ in spin-field-fixed stator coordinates. The rotor flux amount is, as is well known, depending on the required torque, the stator frequency and the DC link voltage set so that the torque is set with minimum stator current and the Asyn-10 chronmaschine 8 is magnetically not saturated. As the stator frequency increases, the required stator flux step increases, so that when the drive limit of the pulse pattern (ctmax) is reached, the flux amount must be reduced in proportion to the reciprocal of the stator frequency, see below. This procedure is state of the art. In a simple way, a memory can be provided in the flow integrator 306, in which the be-calculated rotor flux ψΛ (»+1) is stored. This value can then be used as the current rotor flux ψΛ (») in the next sampling step. A moment calculator 302 calculates according to Eq. (15) for the current sampling step nTa from the predetermined torque setpoint msoii and the rotor flux amount | ψΒ | the torque-forming leakage flux component Δψν in spin-field-fixed coordinates. 20 The leakage flux specification (Δψχ, Δψν) can also be limited by a current limiting device 303 such that a predetermined stator current limit Isma * is not exceeded. A downstream tilt-limiting device 304 can additionally limit the leakage flux specification in such a way that the tilting slip is not exceeded.
    A current limiting and a tilt limitation in the regulation of an asynchronous machine 25 are known per se.
    In a pulse pattern selection unit 305, from the leakage flux vector Δψ -,,, calculated in the tilt limitation device 304, the rotor flux magnitude | ψ "| and e.g. detected with the rotary encoder 11 «! mechanical angular frequency wm the stator loop frequency ω5 calculated.
    With the stator circuit frequency oos, the measured value of the intermediate circuit voltage ud and the stator flux linkage ys determined by the flux integrator 306, as is known, a suitable pulse pattern PM with the modulation limit (½) is selected and the sampling time Ta is determined for the next calculation step selected from the prior art pulse pattern PM, which depends on the stator ring frequency cus and the maximum -10-
    VT-3478 AT permissible switching frequency of the inverter 7 is synchronized with the stator flux linkage ψδ, or asynchronously to constant switching frequency switches.
    The flux integrator 306 calculates from the intermediate circuit voltage ud, the mechanical angular frequency ω, "the leakage flux vector Δψ-, ΧΪ formed by the tilt limiting device 304, and the output limit amai (sampling time Ta, stator circuit frequency ios) formed by the pulse pattern selection device 305, the stator flux step δψ8 Rotor flux magnitude | ψκ | and stator flux linkage ψ8 ·
    A pulse pattern generator 307 now converts the stator flux step δψδ with the measured value of the intermediate circuit voltage uä and the pulse pattern PM selected in the pulse pattern selection 305 into the three control signals S 1, S 2 and S 3 of the inverter 7 using a method known from the prior art.
    Since the intermediate circuit voltage Ud available in the intermediate circuit of the inverter 7 determines the maximum possible stator flux step 5ySmax, advantageously a step limitation can also be provided. As is known, the largest change in the stator flux space vector in terms of magnitude can be achieved if one of the six possible stator voltage space phasors is applied unchanged during the entire sample interval, as is the case with full block clocking. The amount of this maximum possible, can be applied at full block timing voltage time surface is
    i &quot; L usc / t max.lfi = ~ u / ^ i Depending on the pulse method, the semiconductor switch in the inverter 7 is in consideration of the minimum pulse times
    Only a certain proportion of it can be used
    Jusi nr "
    max, MW =, where the excursion limit aDW »Ta i J" fusi / T max.PAf Just / T »L ηαχ, Γ® the ratio of the maximum achievable with a pulse pattern
    Specifies voltage time surface to the maximum achievable during full-cycle clocking voltage time surface. From the above equations can then be given given, limited
    Statorflussschritt S | / S) bg (| öyS bg | = δψ5 ηωχ) calculate the allowable stray flux vector Δψ ^, and a limited rotor flux ψΚ ίκ and a limited stator flux with the allowable leakage flux vector Δψ ^, and thus realize a step limitation Since on the one hand the required size of the leakage flux step results only in the flux integrator 306 and on the other hand the new rotor flux ψκ (// Γ + Ta) is only reached if the required stator flux step 6ψδ with the selected pulse pattern also is executable (δψ5 <δψδ ^), the limitation of the stator flux step size is preferably already taken into account in the flow integrator 306.
    The calculations in the flow integrator 306 could proceed as follows. From the sampling time Ta, the stator circuit frequency cos, the mechanical angular frequency u) m and the parameters of the machine code, the various matrices are calculated. From the rotor flux ψΛ (//) present in the memory, the rotor flux magnitude | ψκ | calculated. The field-fixed scattering flux vector Δψ-xy is transformed into the fixed flux vector Δψ., ΑΡ, which is fixed in the stator. through the unit vector e ^ p in the direction of the rotor flux. Thus, the rotor flux vector ψβ (η + 1) of the next sampling step (n + 1) can be calculated from the leakage flux vector Δψ. ^ Ρ and the, e.g. calculated by the memory, current rotor flux vector | / R (n) are calculated. Thus, from the leakage flux vector Δψ., Αρ and the determined rotor flux vector ψκ (η + 1), the stator flux linkage ψ8 (η + 1) of the next sampling step (n + 1), as well as with, e.g. from the memory provided current Statorflussverkettung ψ $ (η), which are determined in the next sampling step (n + 1) stator flux step δψδ (η + 1) are determined.
    An optional step-limiting device determines from the stator flux linkage ψ5 (η + 1), the rotor flux yR (n + 1) and the desired stator flux step 20 δψδ (η + 1) a limited stator flux step 5vj / s, bg (n + 1), which is from Pulse pattern generator taking into account the intermediate circuit voltage ud, the sampling time Ta and the driving limit amax in the next sampling step (n + 1) is executed. Due to the limitation of the stator flux step 5 | rS) be (n + 1), the calculated rotor and stator flux linkages are not reached in the sampling step (n + 1), therefore starting from 6 | / sibg (n + 1) on the limited 25 Leakage flux vector Δψ ^ (// + 1) is recalculated. With this limited stray flux vector Δψ ^ (// + 1) the rotor and stator flux linkages are calculated, which result with effective limitation for the sampling step (n + 1).
    A feedforward control requires a machine model that is as exact as possible. For this purpose, it is also possible for the parameters of the machine model to be updated online during operation. For this purpose, a parameter adaptation unit 308 is provided which comprises the measured stator current space vector ismess, the stator frequency ωδ, the mechanical angular frequency ojm and the intermediate circuit voltage ud and control signals S /, S2 and S3 formed Statorspannungsraumzeiger the machine parameters identified and the model parameters -12-
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    10 2012/50200 VT-3478 AT
    Stator resistance rs, rotor resistance rR, stator inductance 1 $ and stray inductance! "Depending on the operating state of the asynchronous machine 8. That is, the parameters for the next sampling step are recalculated. In this case, methods known per se from the state of the art are used which are not described in more detail here. In general, this is not necessary, but it is sufficient to track the parameters in a slower task, since the operating state of the asynchronous machine 8 on which the parameters depend (such as temperature, flow) does not change as fast as the torque dynamics. The parameter adaptation can thus be calculated in a slower task than the machine control. The control 9 is preferably in a unit, e.g. a microprocessor, a DSP, a programmable logic controller, etc. implemented.
    An exemplary current limiting device 303 will be explained in more detail with reference to FIG. For current limitation, approximately the proportion of current due to the iron losses is neglected (/ s "is., [Fe ^ 0) and the stator current iss according to Eq. (14) expressed by the 15 flux linkages. If one continues to use fürH + Ay / for the stator flux linkage, the leakage flux linkage can be determined by the stator current and the
    Express the rotor flux to Äy / = --- (/^./ s -ψ). At the current limit [s = with 4- + K ~ 20 ψ e [Ο, 2π the leakage flux passes through a circle whose center on the x-axis is at - Ψ ft | υη ^ has a radius of ISIW [. The stator current thus remains smaller than / Smax as long as the leakage flux pointer Ay remains within this circle. The field-forming component y / x of the leakage flux can therefore be limited to the interval 'Λ, nax - | y /; J &lt; Ay /, &lt; j ^ ~ (lsISmax - | y ^ Ä |). The
    'S + Ισ h + V torque-forming component Ay / ^ .. can be related to the interval - Ay / vnüK <Ay / v <Αψχααοι
    Us νΛ + L X (/ | Λ fSJKBl "Ay / ,. h ---- ly / l I are bounded if the root-J 1 h + L) 25 gument is positive. For Ay /, + 777- ψ &gt; 7 ^ - / Smax Ay / V = 0 can be set. From this, in an arithmetic circuit 401 (e.g., a microprocessor, a DSP, a programmable logic controller, etc.), the stator current limit Ismax is multiplied by the factor / S / CT / (/ 5 + / σ) and the rotor flux amount | ψκ | multiplied by the factor Ισ / (Is + L). With these auxiliary variables at the output of arithmetic circuit 401, in the arithmetic circuit 402, the interval -13-
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    10 2012/50200 VT-3478 AT limits calculated for Δψχ with which in the limiting device 403 the field-forming leakage flux component Δψχ is limited. In the arithmetic circuit 404, the interval limits for Ai //, with which the torque-forming stray flux component Δψγ is limited in the limiting device 405, are calculated. 5 10
    Furthermore, an exemplary tilt limitation will be described with reference to the block diagram in FIG. If the asynchronous machine 8 is operated at the drive limit of the inverter 7, the amount of the stator flux can not be further increased. An increase of the torque-forming stator flux component ψ5}. - Δψ}. is just more up
    Cost of the field-forming stator flux component y / ft = | y / ft | + Ay / v possible. For stationary field conditions Δψχ = 0, y / Sx = ψΗ and the stator flux amount frozen by the intermediate circuit voltage ud and the selected pulse pattern PM at the drive limit
    It was achieved
    | Y / s | The result for the torque m - η-Δψy y / J nen largest possible value thus dm dWSy
    = 0, ie at ij / s = or kiV2 y / i;) = i // &amp; With increasing torque-forming stator flux component y / s &gt;. If the torque increases, the field-forming stator flux component y / Sx must be reduced because of the output limiter. With the tilt limit yjSy = y / a, the torque reaches its maximum value with the so-called tilting moment. If the rotor slip is to be further increased, the field-forming stator flux component y / &amp; smaller than the torque-forming Statorflusskomponente y / Sjl and the torque drops despite slip-20 increase again below the overturning moment. The tilt limitation must therefore satisfy the condition y / ftl &lt; y / Sx, which in steady-state mode is equivalent to limiting the torque-forming leakage flux component Δψγ to ~ ^ α ^ / Λ <Δψ <kKjpp | y / ft | with 0 &lt; kapp &lt; 1 is. This is realized in the Kippbegrenzungseinrichtung 304. In the arithmetic circuit 408, from the rotor flux amount | ψκ | calculates the interval limits with which the torque-forming stray flux component Δψγ is limited in the limiting device 409. -14- E014.1
    Printed: 25-05-2012 10 2012/50200
    VT-3478 AT
    With the limited stray flux components at the outputs of the limiting devices 403 and 409, the limited leakage flux vector Δψ. | Χν, 0β = (Δψ ^ Δψϊιί> 0) Ί is available for further processing.
    -15-
    权利要求:
    Claims (5)
    [1]
    Printed: 25-05-2012 E014.1 H U &quot; 10 2012/50200 VT-3478 AT Claims 1. Method for controlling the torque of an asynchronous machine, characterized in that in stator-fixed coordinates (α, β) starting from the current rotor flux 5 {ψ "(ττ)) and stator flux (ψ5 («) ) the stator flow step to be executed in the next sampling step (n + 1) (δψ5 («+ ΐ)) for achieving a setpoint torque, and the rotor flux (yR (// + l)) and stator flux (ψ8 (η + ΐ)) for the next sampling step (n + 1).
    [2]
    2. The method according to claim 1, characterized in that oriented on the current rotor flux (ψ "(η)) a leakage flux vector (Δψ) with a field-forming leakage flux component 10 (Δι // Λ.) In the direction of the current rotor flux and a torque-forming scattering flux component ( ) is calculated normally standing on the direction of the current rotor flux.
    [3]
    3. The method according to claim 2, characterized in that the leakage flux vector (Δψ) is limited in order to comply with the largest possible, executable Statorflussschritt (δψ8 (/ ι + ΐ)) 15.
    [4]
    4. Method according to claim 1, characterized in that the stray flux vector (Δψ) is limited in a current limiting device so that a predetermined stator current limit is maintained.
    [5]
    5. The method according to any one of claims 1 to 4, characterized in that the leakage flux vector (Δψ) is limited in a Kippbegrenzungseinrichtung, so that a predetermined Kippgrenze is met. -16-
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    引用文献:
    公开号 | 申请日 | 公开日 | 申请人 | 专利标题
    DE102013212054A1|2013-06-25|2015-01-08|Robert Bosch Gmbh|Method and device for operating an asynchronous machine, asynchronous machine|
    WO2014187751A3|2013-05-21|2015-04-09|Robert Bosch Gmbh|Simulation of a field-oriented stator voltage of a stator of an asynchronous machine steadily required during operation|FI79002C|1988-02-08|1989-10-10|Abb Stroemberg Drives Oy|FOERFARANDE FOER MOMENTKONTROLL AV EN VAEXELSTROEMSMASKIN.|
    FI87501C|1990-06-12|1993-01-11|Kone Oy|Procedure for controlling an asynchronous motor|
    FI90163C|1991-10-25|1993-12-27|Abb Stroemberg Drives Oy|Method for determining stator flow in an asynchronous machine|
    WO2008047438A1|2006-10-19|2008-04-24|Mitsubishi Electric Corporation|Vector controller of permanent magnet synchronous motor|
    DK2456064T3|2010-11-17|2015-11-30|Abb Technology Oy|The control method for doubly fed electric generator|
    CN102340278A|2011-09-30|2012-02-01|哈尔滨工业大学（威海）|Method for estimating stator flux of motor in vector converter|
    CN102364871B|2011-10-24|2013-06-05|洛阳理工学院|Method for directly controlling torque of induction motor and control device|
    法律状态:
    优先权:
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    ATA50200/2012A|AT511134B1|2012-05-24|2012-05-24|Method for controlling the torque of an asynchronous machine|ATA50200/2012A| AT511134B1|2012-05-24|2012-05-24|Method for controlling the torque of an asynchronous machine|
    CN201380027136.5A| CN104335480B|2012-05-24|2013-05-22|For the method for the torque for adjusting asynchronous machine|
    EP13725131.0A| EP2856633B1|2012-05-24|2013-05-22|Method for controlling the torque of an induction machine|
    PCT/EP2013/060436| WO2013174827A1|2012-05-24|2013-05-22|Method for controlling the torque of an asynchronous machine|
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